Device for, in particular bistatic radar applications

ABSTRACT

In a device for bistatic radar applications, at least two spaced-apart radar sensors having separate carrier-frequency oscillators are provided, which do not require phase synchronization. The pulse modulation is carried out time-synchronously for all transmitter and receiver pairs. The cross-echo signals can be analyzed in an analyzing unit, in which a mixing of the transmitted and received signals takes place.

FIELD OF THE INVENTION

The present invention is directed to a device for bistatic radarapplications.

BACKGROUND INFORMATION

The fundamental operation of pulsed radars for measuring the distanceand velocity of objects had already been described in 1938 by Col.William Blair of the U.S. Signal Corps. When a microwave carrierundergoes pulse modulation, a signal of a defined pulse duration Tp isperiodically transmitted at the pulse repetition frequency PRF. Thesignal reflected off of an object is attenuated in the receiver to thebaseband range. By analyzing the baseband signal, the signal propagationtime T and possibly the signal Doppler shift f_(D) are determined. Fromthe propagation time T, the object-sensor slant distance R is derived,ultimately based on the speed of light c, from the relationship R=c·T/2,and the object velocity v is determined, with the carrier frequencybeing f_(C), from the relationship v=c/2·f_(D)/f_(C).

Conventional pulsed radar systems use the following operating modes:

LPRF (Low Pulse Repetition Frequency):

In this case, pulsed radars have such a low pulse repetition frequencyPRF, that a unique measurement up to the greatest desired objectdistance is always possible. However, if velocities occur that canresult in f_(D) being greater than PRF/2, the velocity determination isno longer unique.

HPRF (High Pulse Repetition Frequency):

Here, operation takes place at such a high pulse repetition frequencythat the velocity determination in the entire relative velocity range isalways unique. The distance measurement is only unique when all objectsin the detecting range exclusively have smaller distances than c/(2 PRF)to the sensors.

PRF Staggering (Staggered PRF):

To avoid so-called blind speeds which occur at constant pulse repetitionfrequency, or to flatten the line spectrum of the transmitted signalthat exists given a constant pulse repetition frequency, e.g., forimproved interference suppression, pulse pause intervals of variablelength (variable interpulse period VIP) are also used.

Coherent Mixing:

To attenuate the received signal to the baseband range, it is customaryfor the receiver to mix the received signal with a copy of thetransmitted signal. Given a spatial proximity of the transmitter andreceiver, the copy can be possibly derived from the same oscillator asthe transmitted signal or from a second oscillator of the receiver'sown. Depending on whether a stochastic relationship exists from pulse topulse, among the phases of the received signal and its copy, one speaksof incoherent or coherent mixing. The coherent mixing affords a preciseDoppler or velocity determination. However, to achieve the desiredcoherence, considerable outlay must be expended to synchronize thephases (e.g., use of lock-pulse methods or digital detectors of thetransmission phase). Incoherent methods are usually called for when novelocity measurement or only an imprecise velocity measurement isrequired.

Monostatic, Bistatic:

If the transmitting and receiving antennas are “distinctly” spatiallydistant from one another, and if the transmitted signal and its copy arederived from different oscillators for mixing purposes, one usuallyspeaks of bistatic radar systems, in contrast to monostatic radarsystems.

Pulse Compressions:

For a pulsed radar to achieve a minimal coverage range, a minimum oftotal energy is required which must be reflected off of an object andintegrated by the receiver. Given a predefined pulse repetitionfrequency, limited peak power output of the transmitter, and limitedpermissible integration time, the energy can only be still increased byprolonging the pulse duration. On the other hand, the correlationduration (width of the autocorrelation function) of a pulse determinesthe attainable resolution of a pulsed radar. By using internal pulsemodulation/coding, also referred to as pulse compression methods, thecorrelation duration (the resolution) and pulse duration (energy andaverage power output and, thus, instrumented coverage range) can betheoretically defined independently of one another. Customarycompression methods are linear or non-linear frequency modulation, aswell as biphase or multiphase modulation.

It is known that varying combinations and hybrid forms of the abovementioned methods are used.

Fields of Application of Pulsed Radars:

Monostatic Pulsed Radars:

In military applications and in civilian air-traffic control, e.g.,monostatic pulsed radars having substantial transmitting power andantenna directivity (beam focusing) are often used for measuring greatdistances and, to some extent, high velocities. Frequently, a range andazimuth scan is carried out, as well as a relatively complex Dopplerprocessing (MTI (moving target indication), MTD (moving targetdetection) process), as well as, typically, pulse coding/pulsecompression, e.g., chirp (dynamic wavelength change) and modulation ofthe pulse repetition frequency (VIP (variable interpulse period),staggered PRF (pulse repetition frequency)).

Bistatic Pulsed Radars:

Bistatic pulsed radars are found in military applications, in astronomyand in meteorology, where the object distances are large and areaccompanied by great transmitter and receiver distances (for example,baselines in the range of hundreds of kilometers). High demands aretypically placed on the components of such bistatic radars, particularlydue to the requisite time synchronization of the sensors (pulsesynchronization for distance measurement, phase synchronization forvelocity measurement (Doppler)) over large spatial distances. Alsoregarded as difficult are the required synchronized alignment of theviewing directions and, in some instances, allowance for platformmovements.

Low-cost Pulsed Radars:

Microwave pulsed radars are increasingly being used in applicationswhere objects are detected at small distances, using low transmittingpower and a wide visual range, and where, additionally, low costs arerequired, such as for door openers, room surveillance, detection ofmotor-vehicle surrounding fields. Often used in this context aremonostatic LPRF (low pulse repetition frequency) methods, incoherentmixing, no pulse compression, or possibly pulse compression includingbiphase modulation. In contrast to military radar systems or air-trafficcontrol radars, for the low-cost pulsed radars, high-quality componentsare rarely used. Rather, oscillators having low frequency stability,mixers and LNAs (low-noise amplifiers) having low bandwidth and highnoise factor are used, for example.

SUMMARY

The present invention renders possible a cross-echo detection anddistance measurement when working with bistatic pulsed radars, i.e.,with spatially separate transmitting and receiving antennas and carrierfrequency oscillators, it being possible for both carrier frequencyoscillators of any one transmitter/receiver pair, in contrast toconventional bistatic systems, to be run in asynchronous operation,i.e., they do not necessarily have to be frequency-synchronized orphase-synchronized.

The device according to the present invention is distinguished fromcustomary bistatic radar applications, in particular by atime-synchronized pulse modulation when working with transmitting andreceiving sensors.

The present invention may be advantageously applied in connection with apulse repetition frequency that is selected in accordance with the lowpulse repetition frequency method, in particular a pulse repetitionfrequency that is selected to be only slowly changeable over time orpiecewise constant over time.

It is thus possible to implement a bistatic pulsed radar in the low costrange as well, using low transmitting power to measure small cross-echodistances (given small baselines), and components that are not of highquality. In pulsed-radar arrays, it may be used to simultaneouslymeasure direct and cross-echo distances. The additional cross-echodistances increase the spatial sampling of the sensor surroundings, maybe used for classifying object contours, and enhance the redundancy ofthe sensor information.

The time-synchronous pulse modulation of the carriers of adjacentsensors and mixing of transmitted and received signals result in “imagesignals” having frequency components below half of the pulse repetitionfrequency PRF. These are referred to in the following as “cross-echoDopplers”. The mid-frequency of such a cross-echo Doppler may beadjusted via the pulse repetition frequency PRF. The power of thecross-echo Doppler supplies a continuous low-frequency signal, therebyrendering possible cross-echo detection and distance measuring.

In addition to the described implementation, a pulse compression may becarried out. A pulse jitter is likewise possible, provided that it isproduced in such a way that it is synchronous for both sensors, and thatthe cross-echo Doppler is still sufficiently band-limited with respectto the downstream analyzing unit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a time-synchronously controlled pulsed-radartransmitter/receiver pair.

FIG. 2 shows a power density spectrum of the mixed, non-pulsed carriersof adjacent sensors.

FIG. 3 shows a power density spectrum of the mixed, pulsed carriers ofadjacent sensors, given a negligible pulse duration.

FIG. 4 shows the power density spectrum of the mixed, pulsed carriers ofadjacent sensors, given a not insignificant pulse duration.

FIG. 5 shows the power density spectrum of a real I(Q) (in-phase andquadrature) signal during cross-echo reception.

FIG. 6 shows a cross-echo Doppler control and analysis in one embodimentof the present invention entailing little cost outlay.

FIG. 7 shows a cross-echo Doppler control and analysis in one embodimentof the present invention entailing medium cost outlay (spectrum analyzerprinciple).

FIG. 8 shows a cross-echo Doppler control and analysis in one embodimentof the present invention entailing substantial cost outlay (MTD (movingtarget detection) principle).

FIG. 9 shows two transmitting and receiving sensors, includingrepresentation of the direct echoes and Doppler cross-echoes, as well asof the cross-echo range scan.

DETAILED DESCRIPTION

FIG. 1 shows two pulsed-radar sensors 11, 12, of which top sensor 11functions as transmitter (Tx), bottom sensor 12 functions as receiver(Rx). Using their respective carrier-frequency oscillators 21, 22, thesensors generate carrier signals x₁ and x₂ having individual carrierfrequencies f_(L01) and f_(L02). These carrier signals are modulated bythe same pulsed source 3 using the 0–1 pulse sequence p, i.e., viamodulators 51, 52, pulses are impressed on the output signals of thecarrier-frequency oscillators. A separate pulsed signal source 3 mayalso be assigned to each of sensors 11, 12. However, this requiressynchronizing these pulsed signal sources among one another. This may beaccomplished either by using a connecting lead, or otherwise byrecovering the transmitter pulse repetition frequency from the receivedsignal and compensating for the phase offset. The phase offset may bedetermined by utilizing redundancy, since, due to the reversibility ofsignal paths (S11

S12), two cross-echo measurements of an object are normally available,as are possibly existing self-generated-echo measurements of an object.For example, assuming: Δ=phase lead angle from pulsed signal source 1 topulsed signal source 2; tofK=cross-echo propagation time from S11 toobject K to S12 or return direction=cross-echo measurement from S11 toS12 relative to pulsed signal source 2; tofK21=cross-echo measurementfrom S12 to S11 pulsed signal source 1; it then holds that tofK=tofK−Δand tofK=tofK21+Δ→Δ=(tofK12−tofK21)/2→tofK=(tofK12+tofK21)/2. The signalradiated by the transmitter, once reflected off of an object andsubsequently to propagation time (time-of-flight tof), is received bythe receiver. Using a delay circuit/delay line 6, the receiver delayspulse sequence p by delay time τ. If adjusted delay τ corresponds topropagation time tof, then in the case that τ=tof, signal m=p·x₁·x₂results at the output of mixer 7, to which, depending on thetime-synchronous pulse modulation, a transmitted signal, on the onehand, and, a received signal, on the other hand, may be supplied.

This (ideal) mixed signal is itself low-pass filtered, for example, inan analyzing unit 4 having a downstream real amplifier 8 and mixer 7.The I-signal is then available at the output of the amplifier or of theimpedance converter and, in the case of a second mixer which works withthe 90° phase-offset carrier, also a Q-signal, for further low-frequencysignal processing. The following describes the spectrum that results forthe I(Q)-signal.

1. The mixture (multiplication) of the non-pulsed carriers, e.g., x₁ andx₂ in FIG. 1, of two adjacent sensors having average differentialfrequency df=f_(L01)−f_(L02), would produce a spectrum havingband-limited components of around df=f_(L01)−f_(L02) and f_(L01)+f_(L02)(FIG. 2). The summed component may be disregarded due to the low-passcharacteristics of mixer 7 and amplifier 8. The width of the remainingspectral component around df is determined by the short-term frequencystability of the carrier-frequency oscillators during the pulseintegration time. It is important that such a band-limited spectrum alsobe produced when working with oscillators that are not frequency- orphase-synchronized.

2. The pulse modulation of product x₁·x₂, which ultimately leads toideal mixed signal m, corresponds to a sampling, the sampling frequencybeing given by adjusted pulse repetition frequency PRF of the pulsegenerator. However, in the spectrum, an ideal sampling (δ sampling)leads to a periodic continuation of the spectrum of the sampled signal.Thus, the spectrum distributed around df is mirrored twice into each offrequency intervals [z·PRF, (z+1)·PRF], z being an integral number (FIG.3). It should be noted that a band-limited signal is always formed inthe frequency range [0, PFR/2], thus even given differential frequencydf, which is substantially greater than pulse repetition frequency PRF(thus given subsampling). In this context, mid-frequency fa of the“image signal” in [0, PFR/2] and differential frequency df areinterrelated, as expressed by the equationdf=n·PRF+−fa  (1)n being ∈N_(O) (an integral submultiple between df and PRF). An idealsampling is provided when the pulse duration is very short as comparedto the shortest period duration of the sampled signal, i.e., Tp<<1/df.If this is not the case, the amplitudes of the repeated spectralcomponents fall off in accordance with an envelope defined by the pulseshape and the not insignificant pulse duration (FIG. 4). In the case ofa square-wave pulse of length Tp, the envelope is, for example, a sinx/xcharacteristic having the first zero position at 1/Tp.

3. The spectrum of the real IQ-signal falls off markedly above thelimiting frequencies of the mixer and amplifier/impedance converter,which are typically substantially lower than differential frequency dfand generally resemble a characteristic shown in FIG. 5. This limitedsignal component, formed by a cross echo in the I(Q)-signal and havingits essential frequency components below PRF/2, is referred to in thefollowing discussion as cross-echo Doppler. A direct echo of anextremely rapidly moving object having corresponding Doppler frequenciesaround f_(D)=df would lead to a similar signal.

4. It should be noted that image frequency fa of the cross-echo Dopplerhaving predefinable pulse repetition frequency PRF (given a slowlychangeable time frequency df) in accordance with the above equation (1)may be adjusted to a desired value. In particular, by selectivelysetting the pulse repetition frequency, it is possible to ensure, on theone hand, that image frequency fa is always below the limiting frequencyof the mixer and amplifier. On the other hand, given parallel receptionof direct echoes of the sensor, it is possible to ensure that imagefrequency fa is always above maximum Doppler frequency f_(Dmax). Thismay be understood as a “frequency-multiplexing” use of the I(Q)-signal,where the direct echoes and cross echoes are in separate frequencyranges.

An important condition for a distinct separation is that the localmixing oscillators be short-term frequency-stable to such an extent thatthe bandwidth of x₁·x₂ is always smaller than PRF/2−f_(Dmax).

5. Submultiple n and image frequency fa characterize the momentarydifferential frequency of a sensor pair for which cross-echo receptionexists. Thus, when working with sensor arrays having more than twosensors, where the differential frequencies of all relevant sensor pairsdeviate significantly from one another, a transmitter identification isalso possible given a parallel reception of a plurality of cross echoes.

The device according to the present invention provides, for example, thefollowing features:

-   -   synchronous pulsed driving (connecting lead, or by recovering        the transmitter pulse repetition frequency from the received        signal and compensating for the phase offset);    -   using the cross-echo Doppler in I, Q-signals or in signals        derived therefrom, below PRF/2;    -   control/regulation of mid-frequency fa of the cross-echo Doppler        by changing the pulse repetition frequency.

From the above-described features, the following advantages areprovided, for example:

-   -   power measurement (or the like, e.g., amplitude, quasi peak,        etc.) of the cross-echo Doppler supplies a continuous        low-frequency signal for cross-echo detection and cross-echo        distance measuring;    -   for digital processing, cost-effective sampling of the        low-frequency power signal possible, using low sampling rates        (determined by object-sensor dynamics and scan rate);    -   cross echoes may be analyzed in parallel to direct echoes, since        the cross-echo Doppler in the I, Q-signal is placed with the        pulse repetition frequency in a separate frequency range        (frequency-multiplexing operation);    -   a costly phase synchronization of the carriers is not necessary,        but a minimum short-term frequency stability (during the pulse        integration time) of the possibly free-running oscillators is        required;    -   no high demands on the bandwidth of the mixers and low-frequency        amplifiers (above selectable image frequency fa);    -   the mid-frequency of the cross-echo Doppler may be kept constant        via the pulse repetition frequency, e.g., in the case of a        drifting carrier-frequency differential (response to temperature        changes, etc.);    -   active suppression of otherwise sporadically occurring        crosstalk, which occurs in sensor arrays in response to        unsynchronized operation using a fixed pulse repetition        frequency when the cross-echo Doppler, e.g., due to temperature        drift of the carrier frequencies, happens to fall within the        frequency range of the direct echoes (0 . . . f_(D));    -   indirect monitoring of the carrier frequencies as diagnostic        function (built-in test);    -   in sensor arrays, a cross-echo transmitter identification is        possible by estimating the carrier-frequency differential on the        basis of identified cross-echo Doppler mid-frequency fa, the        pulse repetition frequency, and integral submultiple n of the        quotient of df and PRF;    -   customary pulse-compression methods may be used; and    -   cost-effective hardware implementation is possible, e.g.,        variant of an embodiment entailing little outlay in accordance        with FIG. 6, i.e., controllable PRF generator having PLL/DDS,        analog bandpass BP and power measurement/half-wave (one-way)        rectifier.

An example embodiment of the device according to the present inventionhas the following requirements:

-   -   synchronous pulsed driving is necessary (connecting lead, or by        recovering the transmitter pulse repetition frequency from the        received signal and compensating for the phase offset using        redundant measurements);    -   for the fa control, the pulse repetition frequency must be        modifiable in small steps. Thus, the steps must be all the        smaller, the greater the ratio df/PRF is, and the smaller the        bandwidth of a bandpass is selected;    -   in the sensor arrays, for each adjacent sensor and I(Q)-signal,        the power of the cross-echo Doppler must be determined.

Different embodiments of the present invention are described in thefollowing sections.

It is assumed that all example variants of the present invention usecustomary pulse-radar heads in accordance with FIG. 1, i.e., eachtransmitter includes at least one carrier-frequency oscillator andmodulator (or fast-action switch) for pulse modulation; each receiverincludes at least one pulse-delay unit, one carrier-frequencyoscillator, one modulator (or fast-action switch) for pulse modulation,and one mixer for attenuating the received signal (FIG. 1). In a sensorarray, each sensor may be composed of a transmitter and receiver havingonly one carrier-frequency oscillator, which feeds the transmitter andreceiver in parallel.

Thus, the device according to the present invention does not require anymodification of customary radar heads, as used for monostatic operation,as well. All embodiments of the present invention have in common apulse-synchronous driving of all sensors, i.e., of all transmitters andreceivers, and a frequency-selective analysis of the I-signals andoptionally of the Q-signals. The embodiments only exhibit differences inthe signal processing of the I(Q)-signals.

Embodiment Entailing Little Outlay (FIG. 6):

I- and optionally Q-signals are filtered by analog bandpass filters 31using constant resonant frequency f_(res). In this context, using thepulse repetition frequency, an analog or digital control 32 ofmid-frequency fa of the cross-echo Doppler ensures that, in the case ofcross-echo reception, maximum power output is always available in thepass range of the bandpass filters. Power estimation y of the cross-echoDoppler is carried out by analog analysis of the bandpass output signal,e.g., simple rectification (half-wave rectifier/square-law detector) andsmoothing. For digital further processing (detection, distancedetermination, e.g., by scan operation, see detector outputs c_(I)(τ),c_(Q)(τ)), a sampling 33 of signal (y) at a low rate is possible.

Embodiment Entailing Medium Outlay (FIG. 7):

Spectrum analyzer principle: I- and optionally Q-signals are mixed(multiplied) with sinusoidal signals by oscillators 42 tuned to amonitoring frequency f_(mon), e.g., direct digital synthesizer DDS. Ifmid-frequency fa of the cross-echo Doppler is close to f_(mon), alow-pass signal is formed, whose power output, subsequent to low-passfiltering 41, may be estimated analogly or digitally, and analyzed as inthe variant entailing little outlay. The advantage of this embodiment isthat mid-frequency fa does not need to be kept constant, rather thatf_(mon) may follow fa. In addition, the entire spectrum from 0 . . .PRF/2 may be monitored for external interference. Moreover, a samplingof a low-pass signal at a low rate is already possible, and thus anarrow digital low-pass filtering and very precise determination ofpower output.

Embodiment Entailing Greater Outlay (FIG. 8):

I- and optionally Q-signals are sampled 61 pulse-synchronously, i.e., ata rate that is equal to the momentary pulse repetition frequency. Adigital filtering by a bandpass filter bank then follows, compareDoppler filter banks typical of MTP radars 62, and a digital poweroutput estimation. This corresponds to an estimation of the I-, Q-powerdensity spectrum in sub-ranges or to the entire spectrum, from 0 . . .PRF/2. To this end, a numerically efficient FFT (fast Fourier transform)may also be used. The advantage of this embodiment is that the poweroutput and mid-frequency of a cross-echo Doppler may be determined veryreliably from the spectrum, even when, initially, there may be no pastinformation available on the mid-frequency (capture or scan mode). Thereis maximum flexibility with regard to the (digital) faspecification/control. In addition, a reliable detection of interferencesignals is possible. In sensor arrays, the cross-echo Dopplers of alladjacent sensors may be monitored simultaneously.

FIG. 9 illustrates two spaced-apart radar sensors 71 and 72, which areeach equipped for transmitting and receiving operation. The directechoes are denoted by 711 and 721. They are reflected off of wall 8.Object 9 in the nearfield of the radar sensors cannot be detected bythese direct echoes. Detection may only be carried out via cross echo92. Cross echo 91 is reflected off of wall 8. In the cross-echo rangescan likewise depicted in FIG. 9, cross-echo Doppler 92 first appears,conditionally upon the shorter propagation time. Cross-echo Doppler 91appears with a delay that is dependent on the propagation time. Theanalysis of the cross-echo Doppler enhances the spatial sampling in theshort range (angular resolution), permits the classification of objectcontours, and increases redundancy, particularly in the distant range.Hence, on the basis of the two measured values of direct echoes 711 and721 and of cross echo 91, it is possible to verify that object 8 isactually one contiguous reflection surface. If direct echoes 711 and 721were to arrive, but not cross echo 91, then it could be a matter of twodifferent objects in the distant range. It should be noted, however,that given a larger object 9, cross echo 91 could be blocked by wall 8.To increase the reliability of redundancy and detection, a cross-echoanalysis of more than two radar sensors (sensor array) is advantageous.

1. A bistatic radar device, comprising: at least two spaced-apartbistatic radar sensors assigned to one another for performing at leastone of a transmitting operation and a receiving operation; anindependent, asynchronous carrier-frequency oscillator and a modulatorassigned to each one of the at least two radar sensors for impressingpulses generated by at least one pulse-signal source onto an outputsignal emitted by at least one of the carrier-frequency oscillators; ananalyzing unit for a cross-echo Doppler signal, the analyzing unithaving a mixing device, at least one of transmitted and received signalsbeing provided as an output signal of the mixing device; and anarrangement for providing a time-synchronous control of the pulses forthe at least two radar sensors.
 2. The device as recited in claim 1,wherein a single common pulse-signal source is provided for the at leasttwo radar sensors assigned to one another.
 3. The device as recited inclaim 1, wherein the time-synchronous control of the pulses isdetermined by recovering the transmitter pulse repetition frequency, andby compensating for a phase offset on the basis of at least one of theredundant cross-echo measurements and self-generated-echo measurements.4. The device as recited in claim 1, further comprising: a delay circuitfor the time-synchronous control in a signal path between the at leastone pulse-signal source and a modulator of one of the two radar sensors,the delay circuit being configured to be adjusted to effect a signaldelay of the pulses of the pulse-signal source in accordance with apropagation time of radiated radar pulses, until subsequent reception ofreflected radar pulses from an object.
 5. The device as recited in claim1, wherein at least one of the transmitted signal, an assigneddirect-echo Doppler signal, and cross-echo Doppler signal is supplied tothe mixing device.
 6. The device as recited in claim 1, wherein theanalyzing unit is adapted to analyze components of the cross-echoDoppler signal that lie at frequencies below a pulse repetitionfrequency.
 7. The device as recited in claim 1, wherein the analyzingunit is adapted to perform an analog power estimation of the cross-echoDoppler signal.
 8. The device as recited in claim 7, wherein for analogpower estimation of the cross-echo Doppler signal, at least one bandpassfilter is provided, and wherein the analog power estimation includespower estimation of an output of the bandpass filter.
 9. The device asrecited in claim 7, wherein the analog power estimation of thecross-echo Doppler signal is performed by mixing with a tunablesinusoidal signal and subsequent low-pass filtering.
 10. The device asrecited in claim 7, further comprising: an arrangement for sampling atleast one of I-received signal and Q-received signal using the pulserepetition frequency; and an arrangement for at least one of digitalfiltering, frequency analysis, and power estimation of the cross-echoDoppler signal.
 11. The device as recited in claim 1, furthercomprising: an arrangement for one of continuously and intermittentlyregulating the mid-frequency of the cross-echo Doppler signal bychanging a pulse repetition frequency.
 12. The device as recited inclaim 11, wherein the mid-frequency of the cross-echo Doppler signal isregulated on the basis of at least one of a power estimation andfrequency estimation of the cross-echo Doppler signal.
 13. The device asrecited in claim 11, wherein, in addition to the regulation of themid-frequency of the cross-echo Doppler signal, a search is performedfor at least one of the first and repeated tracing of the mid-frequencyof the cross-echo Doppler signal.
 14. The device as recited in claim 11,wherein the mid-frequency of the cross-echo Doppler signal is regulatedin such a way to enable a simultaneous analysis of self-generated echoesand cross-echoes.
 15. The device as recited in claim 11, wherein themid-frequency of the cross-echo Doppler signal is regulated in such away that suppresses a cross feed of cross echoes into the Dopplerfrequency range of self-generated echoes.
 16. The device as recited inclaim 1, wherein the cross-echo Doppler signal is provided formonitoring carrier frequencies of the carrier-frequency oscillators as adiagnostic function.
 17. The device as recited in claim 1, wherein across-echo transmitter identification is provided on the basis ofestimated carrier-frequency differentials, the estimatedcarrier-frequency differentials being based on estimations of activecross-echo Doppler mid-frequency, estimations of an integral submultipleof a quotient of a carrier-frequency differential and pulse repetitionfrequency, and an active pulse repetition frequency.
 18. The method asrecited in claim 17, wherein pulse compression and intra-pulse codingare additionally used for enhancing at least one of interferenceimmunity and transmitter identification.
 19. The method as recited inclaim 1, wherein a synchronous pulse jitter is additionally used forboth radar sensors.